1. Field of the Invention
The present invention relates in general to the field of electronics, and more specifically to a method and system for determining and utilizing a switching parameter of a switching power converter to facilitate transitions between discontinuous conduction mode and critical conduction mode.
2. Description of the Related Art
Many electronic systems include circuits, such as switching power converters to provide efficient power conversion from a voltage supply into a regulated output voltage. When converting an alternating current (“AC”) supply voltage into a regulated output voltage, switching power converters utilize a switch that turns ON and OFF multiple times during a single cycle of the AC supply voltage. The ratio of the ON time of the switch to the OFF time correlates to an average power conversion. Improving the energy efficiency of electrical circuits, including switching power converters, continues to be a high priority for many entities including many companies and countries.
Switching power converters generally operate in one of three different modes: discontinuous conduction mode (“DCM”), critical conduction mode (“CRM”), and continuous conduction mode (“CCM”). CCM tends to be used by switching power converters in higher power, e.g. 300+W, applications. In lower power applications, switching power converters tend to use DCM and/or CRM operational modes. A rectified AC supply voltage rises from zero volts (0V) to a peak voltage, returns to 0V, and repeats. The switch in the switching power converter cycles many times during a single cycle of the rectified AC supply voltage. For example, the switching frequency of the switch is often between 20 kHz and 100 kHz while the frequency of a rectified AC supply voltage is generally between 100 Hz and 120 Hz. Due to, for example, higher switching frequencies at lower voltages in CRM, DCM tends to be more efficient than CRM during the lower voltages of a cycle of the rectified AC supply voltage. Due to, for example, higher conduction losses in DCM at higher voltages, CRM tends to be more efficient than DCM at higher voltages of the rectified AC supply voltage cycle.
FIG. 1 depicts an electronic system 100 that operates in dual conduction mode to take advantage of DCM efficiencies at lower supply voltages and CRM at higher supply voltages. The electronic system 100 receives an AC supply voltage VSUPPLY from voltage supply 102. The supply voltage VIN is, for example, a nominally 60 Hz/110 V line voltage in the United States of America or a nominally 50 Hz/220 V line voltage in Europe and the People's Republic of China. An optional dimmer 104 phase cuts leading and/or trailing edges of the supply voltage VSUPPLY. The input voltage VIN represents the supply voltage VSUPPLY in the absence of phase cutting by the dimmer 104 and represents a phase cut voltage if dimmer 104 phase cuts the supply voltage VSUPPLY. A full-bridge diode rectifier 106 rectifies the input voltage VIN and an electromagnetic interference (EMI) filter 108 attenuates high frequency interference of the switching power converter 112 to generate a rectified input voltage VX. The controller 111 generates a switch control signal CS1 to control the switching power converter 112. The control signal CS1 controls the conductivity of field effect transistor (FET) switch 114 to control the primary current iP to meet the power demands of load 116. For an n-channel FET, the FET conducts (i.e. ON) when during a pulse of the switch control signal CS1 and is nonconductive (i.e. OFF) when the pulse of the switch control signal CS1 ends.
FIG. 2 depicts waveforms 200 associated with a switching power converter 112. Referring to FIGS. 1 and 2, when the FET 114 conducts during a primary charging time period T1, the primary current iP ramps up through the primary coil 118 of transformer 120. The dot convention of transformer 120 and the diode 122 prevent flow of the secondary current iS during the period T1. When the controller 111 generates the switch control signal CS1 to turn FET 114 OFF and period T1 thereby ends, the primary current iP falls to 0, and the voltage across the primary coil 118 reverses (also referred to as “flyback”). During the flyback period T2, the secondary current iS quickly rises and then decays. In DCM and CRM, the flyback period T2 ends when the secondary current iS reaches zero. In DCM, the controller 111 waits for an idle period T3 before beginning a new period TT of the switch control signal CS1 with a pulse of switch control signal CS1. In CRM, as soon as the secondary current iS ends, the controller 111 begins a new period TT of the switch control signal and the primary current iP again ramps up until the end of the period T1.
For each period TT of the switch control signal CS1, the primary charging period T1 equals the duration of the pulse of the switch control signal CS1. Since the controller 111 controls the duration of the pulse of the switch control signal CS1, the controller 111 controls the duration of the charging period T1. The duration of the flyback period T2 is a function of several variables such as parasitic resistances on the secondary side of the transformer 120, such as parasitic resistances of the secondary coil 124, diode 122, capacitor 126, and the secondary current iS drawn by the load 116.
FIG. 3 depicts exemplary waveforms 300 associated with the electronic system 100 for two cycles of the rectified input voltage VX, i.e. for two half-line cycles of the supply voltage VSUPPLY. The waveforms 300 includes superimposed waveforms for the rectified input voltage VX, a representative primary current iP, and representative secondary current iS. The waveforms 300 also include a representative depiction of the switch control signal CS1. The term “representative” is used because typically the frequency of the switch control signal CS1 is 20 kHz to 100 kHz and the frequency of the rectified input voltage VX is 100 Hz to 120 Hz. Thus, for clarity only a subset of the pulses of the switch control signal CS1 and resulting primary current iP and secondary current iS waveforms are shown in the waveforms 300.
Referring to FIGS. 1, 2, and 3, the controller 111 controls the switching power converter 112 to provide power factor correction in addition to regulating the primary side current iP. To provide power factor correction, the controller 111 attempts to make the circuit appear resistive to the voltage supply 102 and, thus, create a linear relationship between the primary current iP and the rectified input voltage VX. Thus, as the voltage VX rises at the beginning of the first charging period T1, the controller 111 generates a pulse of the switch control signal CS1. The controller 111 monitors the rectified input voltage VX via feedforward path 128, and the primary current iP rises during the charging period T1 in correlation with a near instantaneous value of the rectified input voltage VX. The controller 111 also monitors the secondary current iS and, thus, the flyback period T2 via feedback path 130 to determine the power demand of load 116 and to regulate the secondary current iS in accordance with the power demand of load 116.
The charging period T1 and the flyback period T2 are shorter nearer to the zero crossings of the rectified input voltage VX. Zero crossings of the rectified input voltage VX occur in FIG. 3 at times t0, t1, and t2. Since the period TT of the switch control signal CS1 during CRM is T1+T2, the switching frequency 1/TT of the FET 114 would be highest for lower values of the rectified input voltage VX. Higher switching frequencies can correspond to higher switching losses and, thus, lower efficiency. The frequency of the switch control signal CS1 during DCM is 1/(T1+T2+T3). Adding the idle time T3 to the period TT of the switch control signal CS1 decreases the frequency and, thus, increases the efficiency of the FET 114 during DCM. As the rectified input voltage VX increases, the flyback period T2 increases. As the flyback period T2 increases, the frequency of the switch control signal CS1 decreases while transferring more energy to the load 116. Since CRM is more efficient at higher voltages, the controller 111 generates the switch control signal CS1 to operate the switching power converter 112 in CRM at higher values of the rectified input voltage VX. The instantaneous pulses of the primary current iP result in an average primary current iP—AVG that is intended to linearly track the rectified input voltage VX.
FIG. 4 depicts a period versus voltage graph 400 that represents DCM and CRM periods TT of switch control signal CS1 relative to the rectified input voltage VX. Referring to FIGS. 1, 3, and 4, the controller 111 includes a DCM/CRM fixed minimum period TT module 132 to determine when to transition between DCM and CRM operation of switching power converter 112. The DCM/CRM fixed minimum period TT module 132 is configured to always enforce a fixed minimum period TTMIN by generating a DCM/CRM transition signal, which controls whether the controller 111 operates the switching power converter 112 in DCM or CRM. The minimum TTMIN determines a transition point between DCM and CRM operation at VX—TRANS. Thus, the controller 111 operates the switching power converter 112 in CRM as long as the period TT is greater than the fixed minimum period TTMIN and otherwise operates the switching power converter 112 in DCM. The fixed minimum period TTMIN is unresponsive to changes in operating parameters of the electronic system 100 but is relatively simple to enforce since the controller 111 determines the charging period T1 and the DCM/CRM fixed minimum period TT module 132 monitors the flyback period T2 via feedback path 130.
It is desirable to improve efficiency of switching power converters.